Ultra compact multi-band transmitter with robust AM-PM distortion self-suppression techniques

ABSTRACT

A communication device includes a power amplifier that generates power signals according to one or more operating bands of communication data, with the amplitude being driven and generated in output stages of the power amplifier. The final stage can include an output passive network that suppresses suppress an amplitude modulation-to-phase modulation (AM-PM) distortion. During a back-off power mode a bias of a capacitive unit of the output power network component can be adjusted to minimize an overall capacitance variation. A output passive network can further generate a flat-phase response between dual resonances of operation.

REFERENCE TO RELATED APPLICATION

This application is a continuation of U.S. application Ser. No.15/877,879 filed on Jan. 23, 2018, which is a continuation of U.S.application Ser. No. 15/068,179 filed on Mar. 11, 2016, the contents ofwhich are incorporated by reference in their entirety.

BACKGROUND

Modern wireless systems utilize multi-band and multi-mode operations tosimultaneously support multiple different communication standards. Theserapidly growing demands have posed tremendous challenges for futureradio frequency (RF) transmitter development and especially poweramplifiers (PA). One popular solution for multi-band PAs is to directlyassemble several single-band PAs either in a chip or on a multiple-chipmodule. This approach, however, can have several drawbacks, such aslarge chip/module area, increased cost, dedicated antenna interface toeach PAs, possible need of off-chip switches and complicated packaging.Tunable passive networks can also be utilized to achieve multi-bandimpedance matching and power combining for RF PAs. Those tunablecomponents often pose a direct trade-off among passive loss andfrequency range and suffer from reliability concerns of tunablecomponents such as varactors and a switch-cap banks.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 illustrates an exemplary communication device comprising at leastan exemplary power amplifier in accordance with various aspectsdescribed;

FIG. 2 illustrates an exemplary drive circuit and power amplifier inaccordance with various aspects described;

FIG. 3 illustrates a power amplifier component in accordance withvarious aspects described;

FIG. 4 illustrates an example graph of compensation and an outputpassive network in accordance with various aspects described;

FIG. 5 illustrates an example output passive network in accordance withvarious aspects described;

FIG. 6 illustrates an example simulation graph of related to an examplepower amplifier in accordance with various aspects described;

FIG. 7 illustrates a flow diagram of an exemplary method in accordancewith various aspects described.

FIG. 8 illustrates an exemplary mobile communication device having apower amplifier component system in accordance with various aspectsdescribed.

DETAILED DESCRIPTION

The present disclosure will now be described with reference to theattached drawing figures, wherein like reference numerals are used torefer to like elements throughout, and wherein the illustratedstructures and devices are not necessarily drawn to scale. As utilizedherein, terms “component,” “system,” “interface,” and the like areintended to refer to a computer-related entity, hardware, software(e.g., in execution), and/or firmware. For example, a component can be aprocessor, a process running on a processor, a controller, an object, anexecutable, a program, a storage device, an electronic circuit and/or acomputer with a processing device. By way of illustration, anapplication running on a server and the server can also be a component.One or more components can reside within a process, and a component canbe localized on one computer and/or distributed between two or morecomputers. A set of elements or a set of other components can bedescribed herein, in which the term “set” can be interpreted as “one ormore.”

Use of the word exemplary is intended to present concepts in a concretefashion. As used in this application, the term “or” is intended to meanan inclusive “or” rather than an exclusive “or”. That is, unlessspecified otherwise, or clear from context, “X employs A or B” isintended to mean any of the natural inclusive permutations. That is, ifX employs A; X employs B; or X employs both A and B, then “X employs Aor B” is satisfied under any of the foregoing instances. In addition,the articles “a” and “an” as used in this application and the appendedclaims should generally be construed to mean “one or more” unlessspecified otherwise or clear from context to be directed to a singularform. Furthermore, to the extent that the terms “including”, “includes”,“having”, “has”, “with”, or variants thereof are used in either thedetailed description and the claims, such terms are intended to beinclusive in a manner similar to the term “comprising”.

In consideration of the above described deficiencies and continuedobjectives, various aspects for a multi-band communication device, suchas a transmitter that can be a highly linear dual-band mixed-signalpolar power amplifier architecture, can offer a fully integratedsingle-chip solution in bulk CMOS technology while eliminating orsignificantly reducing amplitude modulation to phase modulationdistortions being generated in the power amplification stages.Embodiments herein disclose a power amplifier with self-suppression orself-compensation bias scheme techniques for communication/mobiledevices that involve simultaneous dual-band operation, load-pullimpedance matching, parallel power combining and even-order harmonicrejection to maximize power amplifier efficiency with one compacton-chip transformer without any tunable passive elements or switches.

A communication device, for example, can be a transmitter or transceiverof a mobile phone, or other mobile communicating system that can includea power amplifier that self-mitigates phase distortion as part of theamplification modulation, which can be referred to as amplitudemodulation to phase modulation (AM-PM) distortion. AM-PM distortion canrefer to a form of nonlinear phase distortion caused by nonlinearcharacteristics of a circuit component (e.g., a power amplifier) as afunction of the input amplitude. As the input amplitude is modulated,the phase modulation of the output can behave in a nonlinear manner andthen cause increased out-of-band noise, as well as an increase in errorvector magnitude (EVM).

The power amplifier system being disclosed, for example, can operate toutilize the inherent parasitic capacitances of the output stage (outputpassive network) of the power amplifier to minimize the overallcapacitance variation responsible for the phase distortion. For example,a power amplifier with a plurality of unit power amplifier cells canreceive driver signals according to an operating band of input signalsbeing processed, and an output passive network can combine signals fromeach unit power amplifier cell, delivering the power to an antenna orantenna port, and further suppressing an AM-PM distortion by utilizingflat-phase response of a multi-resonance structure.

The unit power amplifier cells can include several capacitive units inparallel, which are absorbed by the output passive network and become apart of output passive network. The capacitors can be integrated as partof the different unit power amplifier cells for capacitances acrossvarious transistor terminals therein. For example, these capacitors canbe across the drain, source and gate terminal of transistors of eachunit power amplifier cells, and can be charged and discharged inresponse to the course of power amplifier. During power back-off mode,certain capacitive cells can be manipulated with a suppression componenthaving transistors that operate at predetermined modes of operation toactivate or deactivate. The parasitic capacitances being generated canthus be further controlled in a way that mitigates the AM-PM distortionat the output of the power amplifier with the output passive network inorder to implement a self-suppression scheme for suppression of inherentparasitic capacitances causing phase distortion.

In addition, the output passive network of the power amplifier can becoupled to or integrated with the power amplifier as a multi-resonancenetwork to perform impedance matching, power combining, even harmonicssuppression and differential to single-ended conversion across a widefrequency range (one octave) with a single transformer footprint.Additional aspects and details of the disclosure are further describedbelow with reference to figures.

Referring to FIG. 1, illustrated is an exemplary communication or mobiledevice 100 comprising a power amplifier in accordance with variousaspects being described. The communication device 100 can comprise amobile or wireless device, for example, and can further include adigital baseband processor 102, an RF frontend 104 and an antenna port108 for connecting to an antenna 106. The device 100 can comprise anexemplary power amplifier 110 as a part of the digital basebandprocessor 102 or the RF frontend 104. The digital baseband processor 102or the RF frontend 104 can comprise such a power amplifier 110 ormultiple power amplifiers operating or coupled in parallel. The RFfrontend 104 can be coupled to the digital baseband processor 102 andthe antenna port 108, which is configurable with the antenna 106.

In one aspect, the power amplifier 110 can operate to provide a powersignal along a transmitter path for transmissions according to variousoperating bands. The power amplifier 110 can operate in multi-band ormulti-mode operations to simultaneously support multiple communicationstandards with various operating bands. Rapidly growing demands haveposed challenges for future radio frequency (RF) transmitterdevelopment, especially power amplifiers. One solution for a multi-bandpower amplifier can be to directly assemble several single-band PAseither in a chip or on a multiple-chip module. However, this canpossibly incur large chip/module area, increased cost, a dedicatedantenna interface to each power amplifier, possibly the need foroff-chip switches or complicated packaging. Additionally, tunablepassive networks can also be utilized to achieve multi-band impedancematching and power combining for RF power amplifiers. However, thesesolutions suffer from the direct trade-off among passive loss and tuningrange as well as the reliability concerns. To address at least some ofthese issues, the power amplifier 110 can comprise a highly lineardual-band mixed-signal polar power amplifier architecture, which offersa fully integrate single-chip solution in bulk CMOS technology accordingto various aspects or embodiments described.

In one example, the power amplifier 110 can comprise a plurality ofpower amplifiers components or unit power amplifier cells, eachconfigured to provide the power signal along the transmitter path (e.g.,path to antenna port 108) based on driver signals being receivedaccording to one or more operating bands or frequencies. The poweramplifier 110 can be further integrated to an output passive networkthat can be a matching network component. This output stage can combinepower signals processed from the different unit power amplifiercells/components of the power amplifier 110. This output passive networkof the power amplifier 110, for example, can further operate to suppressan AM-PM distortion or phase distortion at an output based on or as afunction of the power amplifier 110 operating in a back-off power modeor a saturation power mode. As such, the output passive network of thepower amplifier component 110 can utilize the flat-phase response acrossthe wide frequency range to achieve robust phase response against to thenon-linear output capacitance variation of the power amplifier. Forexample, the power amplifier component 110 can operate to adjust a biasof any number of the unit power amplifier cell components of the poweramplifier 110 to minimize an overall drain capacitance variation ofalong a full operating power range of operation or operational frequencyrange of operation. As such, the output passive network of the poweramplifier 110 can operate as a multi-resonance network to performimpedance matching, power combining, even harmonics suppression anddifferential to single-ended conversion across a wide frequency range(one octave) with a single transformer footprint. The output passivenetwork (matching resonance network component) of the power amplifiercomponent 110 can further provide a flat-phase response across a widebandwidth, so that a change in output capacitance leads to minimumsignal phase change. Thus, the power amplifier 110 can generate anexcellent AM-PM performance across a wide bandwidth.

Referring to FIG. 2, illustrated is an example communication system witha power amplifier (PA) 110′ in accordance with various aspects orembodiments. The PA 110′ (e.g., a CMOS power amplifier or other PA) cancomprise a driver stage component 202, unit PA cell component(s) 204,and an output passive network or an impedance matching network 206 forgenerating a power signal for transmissions involved in multi-band(e.g., uplink and downlink frequency operating bands) and multi-modeoperations concurrently with different communication standards (e.g.,LTE, 3GPP, etc.). The PA 110′ can generate AM-PM self-suppression tooutput signals, which exploits the inherent PA output power stageparasitic capacitance to compensate the non-linear capacitancevariation. The output passive network 206 can further operate as amulti-resonant matching network with flat-phase response is also used tominimize residual AM-PM distortion throughout operation of the PA 110′over the back-off mode of operation and a saturation mode of operation.For example, the PA 110′ can operate to generate an minimized AM-PMdistortion or phase distortion due to the non-linear capacitance of oneor more unit PA cells, such as by mitigating one or more parasiticcapacitances coupled to one or more transistors (e.g., M1-4 as PMOS,NMOS or another transistor type) of each activated unit PA cell 204.

The driver stage 202 comprises one or more driver stage components X1-X2^(n-1) along one or more single or differential drive paths forgenerating driver signals for power amplification at the PA 110′. Thedriver stage components X1-X2 ^(n-1) process electronic signals (e.g.,radio frequency (RF) voltage signals, V_(RF) ⁻ , V_(RF) ₊ , or thelike), and provide a regulated drive signal to the unit PA cellcomponents 204. The driver stage component(s) 202 can each include oneor more comparators or amplifiers 208 and 210 associated with adifferential drive path, respectively. The driver stage 202 can operateto regulate or control the unit PA cell components 204 by providing biassignals or driver signals to gates of the transistors (e.g., M1 and M2)as well provide a voltage bias (Vbias) to the gates of the thick oxidetransistors M3 and M4, for example, according to different modes ofoperation (e.g., a saturation mode, a back-off power mode) and as afunction of one or more different operating bands being processedaccording to the application of the PA 110′. A drive signal can thusmaintain operation of subsequent stages of the PA 110′ according todifferent characteristics of the unit PA cell components 204.

In one example, the unit PA cell components 204 power/driver signalsfrom the driver stage components 202 for operating one or moretransistors M3 and M4 in a back-off mode and a saturation mode. Theback-off mode can refer herein to a decrease in power being supplied orprovided at the unit PA cell components 204 or any group of transistors,such as M3 and M4 together. The saturation mode can refer to an increasein power where the PA components (e.g., each unit PA cell component 204,the PA component 204, transistor M3 or transistor M4) are fullyoperational or powered, such as above a threshold voltage for thickoxide transistors M3 and M4, or the transistors M1 and M4. Although thePA 110′ is illustrated with differential paths, a single transmissionpath can also be envisioned as one or ordinary skill in the art couldappreciate.

The PA 110′ could generate AM-PM distortion or phase distortion as aresult of changes in the amplitude as well as the fluctuation of variouscapacitances of the unit PA cell components 204, such as from thecapacitances Cgs, Cgd, and Cdb. The capacitors Cgs provides acapacitance across the source terminal and the gate terminal, whichreceives driver signals for driving or powering the transistors M3 andM4. The capacitors Cgd of the unit PA cell components 204 provides acapacitance between the gate terminal and drain terminals of transistorsM3 and M4.

The transistors M3 and M4 can comprise thick-oxide transistors that havea thicker oxide layer than the transistors M1 and M2 comprisingthin-oxide transistors having a smaller or thinner oxide layer. For thePA 110′ cascode topology as illustrated in FIG. 2, the capacitors Cgd ofthe thick-oxide transistors M3 and M4 can be the main contributors ofthe AM-PM distortion of the PA 110′ as capacitors Cgd can be morenon-linear with respect to the power/voltage swing level and further aredirectly loaded at the output passive network 206 of the PA 110′. Thisnon-linear capacitance Cgd loading to the output passive network 206 ofPA 110′ can shift the resonance frequency of the output passive network206 of the PA 110′ (resonance frequency can normally be tuned at themaximum power level), resulting in the phase distortion according to theoutput power level (as in AM-PM distortion). The capacitance ofcapacitor Cgd can be related to the width (W) and length (L) of thetransistor device (e.g., M3, M4), the gate-drain overlap capacitance perunit width (Coy) and the total gate capacitance (Cgg).

One way to address the AM-PM phase distortion generated from the poweramplifier component 110′ is to compensate the phase distortion of theunit PA cells 204 at the driver stages 202 using a varactor or capacitorbank based on a look-up table. However, additional memory and processorpower could be utilized, which increases the cost and reduces theoverall power efficiency, which is especially true for widebandmodulated signals (>20 MHz). Therefore, the unit PA cell components 204or output power stage 204 comprises a self-compensating function (e.g.,a suppression component) with respect to the non-linear capacitancevariations, without introducing extra components: as the power isreduced, the bias of the unit PA cells 204 while in a turned off state(back-off mode) can be adjusted to minimize the overall draincapacitance variation.

In one embodiment, as the voltage swing at the drain node of M3 and M4is increasing (PA power increasing), the cascode transistors (M3 and M4)or the thick oxide transistors are operated for a longer time in thetriode region or mode of operation where each capacitor Cgd presents alarger capacitance (W×Cov+W×L×Cgg/2) than the capacitances of each Cgdbeing operated in a saturation region (W×Cov) or mode of operation. Inother words, when the PA 110′ output power is decreasing (back-off modeof operation), the PA 110′ effective capacitance (Cdev) at the drain ofthe cascode transistors (M3 and M4) is decreasing.

In one embodiment, the output power network 204 can self-compensate forthe phase distortion for an effective capacitance reduction at the powerback-off mode of the PA 110′, without additional components, byutilizing the parasitic capacitance Cds of the cascode transistors (M3and M4) via the suppression component 220 comprising transistors M1 andM2, for example. Rather than compensating for the phase distortion ofthe PA 110′ at the driver stages, such as by using a varactor orcapacitor bank together with a look-up table, the unit PA cellcomponents 204 can utilize its own components to self-compensate ormitigate phase distortion. The suppression component utilizes aninherent parasitic capacitance of the power amplifier component 220 toself-compensate a nonlinear capacitance variation at the output.

For example, each of the unit PA cells 204 can include correspondingunit PA cells Y1-Y2 ^(n-1) that can operate in a power-on and apower-off mode depending upon a change in. As such, during operation theunit PA cells Y1-Y2 ^(n-1) with transistors M3 and M4 can fluctuatebetween increasing in power during a power-on phase or mode and apower-off phase or mode of operating. The power-on mode can comprisesaturation mode, for example, in which the PA and any number of outputpower networks Y1-Y2 ^(n-1) of the output power stage 204 are beingfully powered.

Additionally, the power-off phase or mode can be the back-off mode wherepower is being decreased or the output power network of the unit PA cellcomponents 204 is powered down or off. Incidental to this operation,parasitic capacitance is still being generated, but this parasiticcapacitance is not affecting the output because the capacitors Cds canbe effectively floating during the back-off mode of operation since boththe thin oxide transistors M1 and M2 are powered off.

The suppression component 220 can include the transistors M1 and M2, forexample. The suppression component 220 can operate to adjust a bias oftransistors of M1, M2, M3, and M4. The capacitor or capacitive unit ofCds across the drain and source of the thick oxide transistors M3 and M4of each unit PA cell components 204 of Y1-Y2 ^(n-1) are manipulated tominimize an overall capacitance variation in the back-off power mode.The parasitic capacitance of the PA 110′ or one or more unit PA cells204 can have a nonlinear behaviour with respect to a power level wherethe effective parasitic capacitance is decreasing as power isdecreasing. Therefore, the parasitic capacitance varies between thedifferent modes of a back-off mode and a saturation mode of operation.The self-suppression or self-compensation bias scheme generated by thesuppression component can linearize the non-linear parasitic capacitancebehavior of the PA 110′. The suppression component 220 thus enables acontinued base line operation and minimizes an overall capacitancevariation to reduce the phase distortion being generated due to thechanges in a parasitic capacitance between the different modes of normaloperation. Additional details of the operation of the suppressioncomponent 220 are illustrated and described below with reference to FIG.3.

In another embodiment, the output passive network 206 (as an impedancematching network) can be implemented with a single transformer. Thesingle transformer for output passive network 206 include two inductors;one for magnetizing inductance and the other for leakage inductance,parasitic capacitors, and absorb the power amplifier output capacitorsto provide a real impedance transformation or a flat-phase response tothe PA 110′ along a broad bandwidth (e.g., about 2.4 GHz to about 5.5GHz, or other broad band). For example, the output passive network 206can operate as a multi-resonance network to perform impedance matching,power combining, even harmonics suppression and differential tosingle-ended conversion across a wide frequency range (one octave) witha single transformer footprint. The output passive network 206 canfurther generate or provide a flat-phase response across a widebandwidth or at least two different operating frequency bands (e.g.,about 2 GHz and 5.5 GHz), so that a change in output capacitance due tothe non-linear capacitance of the power amplifier leads to minimumsignal phase change. The flat-phase response generated by the outputpassive network 206 can effectively suppress the AM-PM distortion. Thisleads to excellent AM-PM performance across the wide bandwidth range.

An advantage of the PA 110′ is it utilizes the PA transistor's (e.g., M3and M4) inherent parasitic capacitance to self-compensate or linearizethe non-linear capacitance variations, which provides a highly efficientand compact scheme at PA back-off modes of operation among or back forthbetween back-off and saturation mode. Compared to a multi-band PA whichuses individual output matching networks, the proposed multi-band PAoutput stage utilizes only one compact passive transformer as thematching resonance network component 206, which can provide paralleloutput power combining, output impedance matching, even-order harmonicrejection and differential to single-ended conversion across a widebandwidth without any lossy tunable passive elements or switches.Another advantage is that the PA 110′ can significantly reduce thetransmitter area by factor of 2× or more and maximize the PA efficiency.Additionally, for example, the proposed PA 110′ architecture achievesexcellent AM-PM characteristic (<3°), about 30˜40% power addedefficiency (PAE), with 2.05% error vector magnitude (EVM) and 256quadrature amplitude modulation QAM and can cover the wide frequencyrange (1:2 range) with ultra-compact area which is the state-of-artperformance among CMOS PAs.

Referring to FIG. 3, illustrated is an additional example of a unit PAcell component 204 for a PA in accordance with various aspects orembodiments being described. Further, the PA component 204 or PA 110′discussed herein is not limited to digital PAs and can also be used withanalog PAs, or a combination thereof. The example unit PA cellcomponents 204 depicts operation of the output power networks Y1-Y2^(n-1) in two different power stages that can change or alternatinglyoperate during different power levels of operation of the PA 110 or 200.

In one embodiment, the unit PA cell components 204′ can operate togenerate self-compensation or self-suppression of nonlinearities beinggenerated by the parasitic capacitances generated by the differentpowering on and off modes (back-off or saturation modes). For example, afirst power stage comprises the saturation (active) mode of operation302 where the output power networks Y1-Y2 ^(n-1) of the unit PA cellcomponents 204 operate with full or complete power above a thresholdvoltage so that the transistors M3 and M4 of any one of the networksY1-Y2 ^(n-1) are operational, and a channel has been created for currentflow. This allows current to flow between the drain and source. Sincethe drain voltage is higher than the source voltage, the current flow ofelectrons spread out, and conduction is not through a narrow channel butthrough a broader, two- or three-dimensional current distributionextending away from the interface and deeper in the substrate.

In contrast, a back-off mode 304 of operation occurs when the power isdecreased normally and power is backed off so that the transistors M3and M4 are cut-off or in sub-threshold mode. While the current betweendrain and source should ideally be zero when the transistor is beingused as a turned-off switch, there can be a weak-inversion current,sometimes called subthreshold leakage. The subthreshold I-V curve candepend exponentially upon threshold voltage, introducing a strongdependence on any manufacturing variation that affects thresholdvoltage, for example: variations in oxide thickness, junction depth, orbody doping that change the degree of drain-induced barrier lowering.The resulting sensitivity to fabricational variations can complicateoptimization for leakage and performance.

The unit PA cells 204 can comprise n-bit binary weighted power cellswith a differential cascode amplifier topology. A digital switching PAscheme is illustrated in FIG. 3 that operates to turn-on/off the binaryweighted unit power amplifier cells Y1-Y2 ^(n-1) to control theamplitude. For example, when the unit PA cell (Y1) is turned-on insaturation/power mode 302, the cascode transistors (M3 and M4) can bebiased at a high voltage (above a threshold voltage or a saturationpower level) and the thin-gate transistors (M1 and M2) can be drivendifferentially (differential pulse 306 and 308 by the driver stage 202of FIG. 2). When the unit PA cell (Y1) is turned-off in sub-thresholdvoltage or back-off mode 304, the cascode transistors (M3 and M4) can bebiased at a low voltage (below a threshold voltage as LOW<V_(TH)).However, in response to the thin-gate transistors (M1 and M2) beingturned off as well AM-PM distortion can exist.

In one embodiment, the suppression component 220 operates to take intoconsideration the Cds of the cascode transistors (M3 and M4), when theunit PA cells 204 are turned off (or power is decreasing in back-offmode). During back-off mode of operation only a small portion of Cds isloaded at the drain, as the thin-gate transistors are completely turnedoff and one terminal of the Cgd is effectively floating. When the powercell is turned-off or in back-off mode 304, the cascode transistors (M3and M4) can be biased at a low voltage below a threshold voltage (<Vth)to be weekly conducting, while the thin-gate transistors (M1 and M2) asthe suppression component 220 are now completely turned on or fullypowered. Thus, the Cds loads the drain of the cascade transistors (M3and M4) entirely, so that off-state capacitance can be increased. Thecascode transistors (M3 and M4) are thus still biased at sub-thresholdregion (<Vth) to minimize leakage in off state 304.

Consequently, the suppression component 220 can provide additionalcapacitance that can compensate the capacitance reduction of Cgd atpower back-off of PA, without extra control bits when the power cellsare turned-off (when PA power decreases and an effective capacitanceCdev decreases) for a self-suppression or compensation scheme. In otherwords, the total capacitance change resulting in AM-PM distortions isreduced and the unit PA cell compensates or suppresses nonlinearities byadding a parasitic capacitance by coupling the Cds to ground during thepower off or back-off phases of operation. This compensation scheme canbe performed without adding additional capacitors or capacitorcomponents.

In another aspect, the suppression component 220 can comprise adetection component 310 and 312, which can include a set of switches orinverters 312 and 314, configured to detect the back-off power mode ofoperation from the saturation power mode of operation based on a powerlevel of the first plurality of transistors. Based on a detection of thepower level, the inverters 312 or 314 can operate the suppressioncomponent 220 to mitigate the non-linear behavior of the PA component204′ where the effective parasitic capacitance is decreasing as thepower is decreasing and the self-compensating bias operations linearizethe non-linear parasitic capacitance of the PA component 204′.

Referring to FIG. 4, illustrated is an example result of a simulatedself-compensation scheme together with the output passive networkaccording to various aspects being described. The characteristics ofbehavior for the PA 110′ capacitance can be seen from the simulationgraph 400. The curve 402 demonstrates the differential between the totalcapacitance of the PA 110′ during operation while the unit PA cells areoff, low or in back-off, while processing transmission withself-suppression or a compensation scheme via the suppression component220. The curve 404 demonstrates the differential between the totalcapacitance of the PA 110′ during operating while the unit PA cells areoff, or in back-off, while processing transmission withoutself-suppression or a compensation scheme via the suppression component220. As such, FIG. 4 illustrates that the simulated PA 110′ effectivecapacitance (Cdev) at the drain of the cascode transistors (M3 and M4)variation with the suppression components 220 self-compensation schemeis reduced to about 0.4 pF, while without the PA scheme about a 1.31 pFvariation can result with an output power range from 10 mW to 640 mWmwhich is a source or cause of the AM-PM distortion. Therefore, reducingthe total change in capacitance via the suppression component 220self-compensates for the nonlinear capacitance and cuts the change intotal capacitance between operating modes substantially by nearly halfthe total capacitance otherwise during back-off mode. The AM-PMdistortion can be further minimized below 3° for example across afrequency range of 2.4 GHz to 6 GHz, which can cover most of commercialstandard bands.

Referring to FIG. 5, illustrates is an example of the output passivenetwork 206′ in accordance with various aspects being described alongwith FIG. 6 illustrating an output simulation 600 with curvesdemonstrating effects of operations of the output passive network 206′.The output passive network 206′ can operate as a multi-order resonancenetwork that provides to a flat-phase response, in which the slope ofthe phase response is about zero (≈0) within the wide/broad frequencyrange.

The output passive network 206′ can comprise a plurality of inductorsand capacitors configured to provide a real impedance or a flat-phaseresponse to the PA 110′ or the unit PA cells Y1-Y2 ^(n-1) in combinationover a broad/wide bandwidth. The flat phase response is illustrated as asecond phase (2) of the simulation 600. For example, the output passivenetwork 206′ can comprise a second-order resonance network that canresonate at two (or more) different frequencies (e.g. 2 GHz and 5.8GHz). Due to the dual (or multi-) resonance of the output passivenetwork 206′, the flat-phase response is achieved across the frequencyrange within at least the two resonance frequencies. The flat-phaseregion (2) of FIG. 6 indicates a minimal phase variation with respect toloading capacitance variations (Cdev). As such, FIGS. 5 and 6 illustratethe schematic and simulation result of our proposed dual resonance(e.g., resonates at 2 GHz and 5.8 GHz) network clearly showing the flatphase response between about 2 GHz and 5.8 GHz. Other ranges can also beenvisioned as one of ordinary skill in the art could appreciate.

The different inductors Lpx(1−k²) and Lpxk² of the output passivenetwork 206′ can form or be realized as a single transformer 500 for ahigh-order LC matching network that generates a robust phase response orflat-phase response on the nonlinear capacitances of the PA 110′, forexample. The inductors and capacitors of the output passive network canbe realized by the single transformer 500 by utilizing the parasiticcomponents of a physical transformer and the non-linear parasiticcapacitance of the PA component 204, for example.

The single transformer 500 can further operate to efficiently combinethe power from the unit PA cells Y1-Y2 ^(n-1) of the PA component 204 ofFIG. 2, for example. As such, power can be efficiently delivered to anantenna 106 or load, for example, and AM-PM distortion be suppressed atthe same time by the output passive network 206′. Because the PA 110′,for example, provides different capacitance at different power levels,the output passive network 206′ of the PA 110′ can operate to make thephase response of the output passive network robust to the capacitancevariations by utilizing the multi resonance operation over a wideoperation band.

The curve 602 of FIG. 6 can represent an imaginary value or thecapacitive curve. The curve 604 can represent the real value or theinductive curve. The curve 606 can represent the phase response curvewith a flat response in section (2) within a wide band frequencyoperation range, for example. Conventionally, a 1st-order L-C resonancebased output passive network can be widely used for a narrow band poweramplifier. However, the phase response of the 1st-order resonancenetwork can be vulnerable to the loading capacitance variations (Cdev)as it directly shifts the resonance frequency, resulting in undesiredphase shift/distortion. The slope of the phase response thus woulddepend on the loaded Q and the phase distortion due to the Cdevvariation, and can be directly proportional to the slope of the phaseresponse in this case. In contrast, the output passive network 206′generates a flat-phase response across a frequency range within at leasttwo resonance frequencies of different resonance frequencies of signalsof different operating bands. This provides a robust phase response tothe non-linear capacitance variation of the PA 110′.

While the methods described within this disclosure are illustrated inand described herein as a series of acts or events, it will beappreciated that the illustrated ordering of such acts or events are notto be interpreted in a limiting sense. For example, some acts may occurin different orders and/or concurrently with other acts or events apartfrom those illustrated and/or described herein. In addition, not allillustrated acts may be required to implement one or more aspects orembodiments of the description herein. Further, one or more of the actsdepicted herein may be carried out in one or more separate acts and/orphases.

Referring to FIG. 7, illustrated is an example method for utilizing a PAcircuit with AM-PM distortion compensation via an output passive networkfor a communication device (e.g., a mobile device, or user equipment).The method 700 initiates at 702 with providing, via a power amplifier,power signals along a transmitter path.

At 704, the method comprises combining, via an output passive network,the power signals, and suppressing an amplitude modulation-to-phasemodulation (am-pm) distortion based on whether the power amplifier isoperation in a back-off power mode or a saturation power mode. Thecombining, via the output passive network, the power signals comprisescombining the power signals from unit power amplifier cells andproviding an optimum impedance to the power amplifier with a broadbandwidth via a single transformer.

The method can further include adjusting a bias of a capacitive unit ina unit power amplifier cell of the power amplifier to minimize anoverall capacitance variation in the back-off power mode by utilizing aninherent parasitic capacitance of the power amplifier component toself-compensate a nonlinear capacitance variation at the output.

The method can include the output passive network 206 generating aflat-phase response across a frequency range within at least tworesonance frequencies of the different resonance frequencies of theoutput passive network.

Providing the power signal along the transmitter path can furthercomprise operating in the back-off power mode and the saturation powermode alternatingly or sequentially, and in the back-off power modeincreasing a capacitance of a first plurality of transistors atrespective drain terminals, while decreasing the capacitance (e.g., aneffective or differential capacitance) in the saturation power mode. Asecond plurality of transistors, having a smaller gate than the firstplurality of transistors can be provided and coupled at the drains ofthe first plurality of transistors, while the first plurality oftransistors operate in a below threshold voltage region.

To provide further context for various aspects of the disclosed subjectmatter, FIG. 8 illustrates a block diagram of an embodiment of accessequipment, user equipment (e.g., a mobile device, communication device,personal digital assistant, etc.) or software 800 related to access of anetwork (e.g., base station, wireless access point, femtocell accesspoint, and so forth) that can enable and/or exploit features or aspectsof the disclosed aspects.

The user equipment or mobile communication device 800 can be utilizedwith one or more aspects of the converter systems or devices describedaccording to various aspects herein. The mobile communication device800, for example, comprises a digital baseband processor 802 that can becoupled to a data store or memory 803, a front end 804 (e.g., an RFfront end, an acoustic front end, or the other like front end) and aplurality of antenna ports 807 for connecting to a plurality of antennas806 ₁ to 806 _(k) (k being a positive integer). The antennas 806 ₁ to806 _(k) can receive and transmit signals to and from one or morewireless devices such as access points, access terminals, wirelessports, routers and so forth, which can operate within a radio accessnetwork or other communication network generated via a network device(not shown). The user equipment 800 can be a radio frequency (RF) devicefor communicating RF signals, an acoustic device for communicatingacoustic signals, or any other signal communication device, such as acomputer, a personal digital assistant, a mobile phone or smart phone, atablet PC, a modem, a notebook, a router, a switch, a repeater, a PC,network device, base station or a like device that can operate tocommunicate with a network or other device according to one or moredifferent communication protocols or standards.

The front end 804 can include a communication platform, which compriseselectronic components and associated circuitry that provide forprocessing, manipulation or shaping of the received or transmittedsignals via one or more receivers or transmitters 808, a mux/demuxcomponent 812, and a mod/demod component 814. The front end 804, forexample, is coupled to the digital baseband processor 802 and the set ofantenna ports 807, in which the set of antennas 806 ₁ to 806 _(k) can bepart of the front end. In one aspect, the mobile communication device800 can comprise a PA component/system 810 according toembodiments/aspects described herein.

The user equipment device 800 can also include a processor 802 or acontroller that can operate to provide or control one or more componentsof the mobile device 800. For example, the processor 802 can conferfunctionality, at least in part, to substantially any electroniccomponent within the mobile communication device 800, in accordance withaspects of the disclosure. As an example, the processor can beconfigured to execute, at least in part, executable instructions thatcontrol various modes or components of the PA component/system 810(e.g., the system 110, 200, 110, 110′, or 204).

The processor 802 can operate to enable the mobile communication device800 to process data (e.g., symbols, bits, or chips) formultiplexing/demultiplexing with the mux/demux component 812, ormodulation/demodulation via the mod/demod component 814, such asimplementing direct and inverse fast Fourier transforms, selection ofmodulation rates, selection of data packet formats, inter-packet times,etc. Memory 803 can store data structures (e.g., metadata), codestructure(s) (e.g., modules, objects, classes, procedures, or the like)or instructions, network or device information such as policies andspecifications, attachment protocols, code sequences for scrambling,spreading and pilot (e.g., reference signal(s)) transmission, frequencyoffsets, cell IDs, and other data for detecting and identifying variouscharacteristics related to RF input signals, a power output or othersignal components during power generation.

The processor 802 is functionally and/or communicatively coupled (e.g.,through a memory bus) to memory 803 in order to store or retrieveinformation necessary to operate and confer functionality, at least inpart, to communication platform or front end 804, the PAcomponent/system 810 and substantially any other operational aspectsdescribed herein.

Examples herein can include subject matter such as a method, means forperforming acts or blocks of the method, at least one machine-readablemedium including executable instructions that, when performed by amachine (e.g., a processor with memory or the like) cause the machine toperform acts of the method or of an apparatus or system for concurrentcommunication using multiple communication technologies according toembodiments and examples described.

Example 1 is a communication system comprising: a power amplifiercomprising a plurality of unit power amplifier cells configured toprovide a power signal along a transmitter path; and an output passivenetwork component configured to combine power signals from the pluralityof unit power amplifier cells and suppress an amplitudemodulation-to-phase modulation (AM-PM) distortion.

Example 2 includes the subject matter of Example 1, further comprising:a detection component configured to detect a back-off power mode from asaturation power mode based on a power level of the output.

Example 3 includes the subject matter of any of Examples 1-2, includingor omitting any elements, further comprising: a suppression componentconfigured to adjust a bias of a unit power amplifier cell of the poweramplifier to minimize an overall capacitance variation in a fulloperating power range.

Example 4 includes the subject matter of any of Examples 1-3, includingor omitting any elements, wherein the suppression component is furtherconfigured to utilize an inherent parasitic capacitance of the poweramplifier to self-compensate a nonlinear capacitance variation at theoutput.

Example 5 includes the subject matter of any of Examples 1-4, includingor omitting any elements, wherein the output passive network componentcomprises a matching network configured to generate an output impedancematching operation with a single transformer by using parasiticcomponents of the single transformer to generate broadband impedancetransformation and suppress the AM-PM distortion.

Example 6 includes the subject matter of any of Examples 1-5, includingor omitting any elements, wherein the output passive network componentcomprises a plurality of capacitors and an inductor configured toprovide real impedance with a flat-phase response to the power amplifieralong a broad bandwidth to suppress the AM-PM distortion.

Example 7 includes the subject matter of any of Examples 1-6, includingor omitting any elements, wherein the plurality of capacitors and theinductor of the output passive network component are realized by asingle transformer by utilizing parasitic components of the singletransformer and a non-linear parasitic capacitance of the poweramplifier.

Example 8 includes the subject matter of any of Examples 1-7, includingor omitting any elements, wherein the matching network is furtherconfigured to generate a flat-phase response across a frequency rangewithin at least two resonance frequencies of different resonancefrequencies of the matching network, wherein the matching networkcomprises a robust phase response to a non-linear capacitance variationof the power amplifier.

Example 9 is a mobile communication device comprising: a power amplifiercomprising a plurality of unit power amplifier cells, configured toprovide a power signal to a signal processing path, comprising an outputpower stage; and an output passive network of the output power stageconfigured to combine the power signal and suppress an amplitudemodulation-phase modulation (AM-PM) distortion in a back-off power modeof operation from a saturation power mode of operation.

Example 10 includes the subject matter of any of Example 9, including oromitting any elements, wherein the plurality of unit power amplifiercells provide power signals for a desired power to the output passivenetwork, wherein the power amplifier comprises a parasitic capacitancethat has a non-linear behavior with respect to a power level, whereinthe parasitic capacitance decreases as power is decreasing and asuppression component is configured to provide a self-compensating biasscheme that linearizes the parasitic capacitance of the power amplifier.

Example 11 includes the subject matter of any of Examples 9-10,including or omitting any elements, further comprising: a suppressioncomponent configured to adjust a bias of a capacitive unit in a unitpower amplifier cell of the plurality of unit power amplifier cells tolinearize a non-linear behavior of a parasitic capacitance of the unitpower amplifier cell.

Example 12 includes the subject matter of any of Examples 9-11,including or omitting any elements, wherein the suppression component isfurther configured to compensate for an effective capacitance reductionin the back-off power mode of operation of the power amplifier by usinga parasitic capacitance of a first plurality of transistors via a secondplurality of transistors coupled to drain terminals of the firstplurality of transistors.

Example 13 includes the subject matter of any of Examples 9-12,including or omitting any elements, wherein in a back-off power mode ofoperation, a first plurality of transistors of the power amplifier isconfigured to comprise a lower voltage and, in the saturation power modeof operation, the first plurality of transistors comprises a saturationvoltage.

Example 14 includes the subject matter of any of Examples 9-13,including or omitting any elements, further comprising: a suppressioncomponent comprising a second plurality of transistors, coupled to thefirst plurality of transistors, configured to be fully powered in theback-off power mode of operation; and a detection component, comprisinga set of switches, configured to detect the back-off power mode ofoperation from the saturation power mode of operation based on a powerlevel of the first plurality of transistors.

Example 15 includes the subject matter of any of Examples 9-14,including or omitting any elements, wherein the suppression component isconfigured to bias the first plurality of transistors of the outputpower stage while operating below an operational voltage thresholdduring the back-off power mode of operation.

Example 16 includes the subject matter of any of Examples 9-15,including or omitting any elements, wherein the output passive networkcomponent comprises a matching network configured to generate an outputimpedance matching operation with a single transformer by usingparasitic components of the single transformer to generate broadbandimpedance transformation and suppress the AM-PM distortion.

Example 17 includes the subject matter of any of Examples 9-16,including or omitting any elements, wherein the matching network isfurther configured to resonate at different resonance frequencies andgenerate a flat phase response across a frequency range within at leasttwo resonance frequencies of the different resonance frequencies.

Example 18 includes the subject matter of any of Examples 9-17,including or omitting any elements, wherein the power amplifiercomprises the plurality of unit power amplifier cells configured toprovide a power signal along the signal processing path according todifferent operating bands.

Example 19 is a method for a communication system comprising: providing,via a power amplifier, power signals along a transmitter path; andcombining, via an output passive network, the power signals, andsuppressing an amplitude modulation-to-phase modulation (AM-PM)distortion based on whether the power amplifier is operation in aback-off power mode or a saturation power mode.

Example 20 includes the subject matter of Examples 19, including oromitting any elements, further comprising: adjusting a bias of acapacitive unit in a unit power amplifier cell of the power amplifier tominimize an overall capacitance variation in the back-off power mode byutilizing an inherent parasitic capacitance of the power amplifiercomponent to self-compensate a nonlinear capacitance variation at theoutput.

Example 21 includes the subject matter of any of Examples 19-20,including or omitting any elements, wherein the combining, via theoutput passive network, the power signals comprises combining the powersignals from unit power amplifier cells and providing an optimumimpedance to the power amplifier with a broad bandwidth via a singletransformer.

Example 22 includes the subject matter of any of Examples 19-21,including or omitting any elements, further comprising: generating aflat-phase response across a frequency range within at least tworesonance frequencies of the different resonance frequencies of theoutput passive network.

Example 23 includes the subject matter of any of Examples 19-22,including or omitting any elements, wherein providing the power signalalong the transmitter path comprises operating in the back-off powermode and the saturation power mode alternatingly, and in the back-offpower mode increasing a capacitance of a first plurality of transistorsat respective drain terminals, and decreasing the capacitance in thesaturation power mode.

Example 24 includes the subject matter of any of Examples 19-23,including or omitting any elements, further comprising: powering asecond plurality of transistors, having a smaller gate than the firstplurality of transistors and coupled at the drains of the firstplurality of transistors, while the first plurality of transistorsoperate in a below threshold voltage region.

The above description of illustrated embodiments of the subjectdisclosure, including what is described in the Abstract, is not intendedto be exhaustive or to limit the disclosed embodiments to the preciseforms disclosed. While specific embodiments and examples are describedherein for illustrative purposes, various modifications are possiblethat are considered within the scope of such embodiments and examples,as those skilled in the relevant art can recognize.

In this regard, while the disclosed subject matter has been described inconnection with various embodiments and corresponding Figures, whereapplicable, it is to be understood that other similar embodiments can beused or modifications and additions can be made to the describedembodiments for performing the same, similar, alternative, or substitutefunction of the disclosed subject matter without deviating therefrom.Therefore, the disclosed subject matter should not be limited to anysingle embodiment described herein, but rather should be construed inbreadth and scope in accordance with the appended claims below.

In particular regard to the various functions performed by the abovedescribed components or structures (assemblies, devices, circuits,systems, etc.), the terms (including a reference to a “means”) used todescribe such components are intended to correspond, unless otherwiseindicated, to any component or structure which performs the specifiedfunction of the described component (e.g., that is functionallyequivalent), even though not structurally equivalent to the disclosedstructure which performs the function in the herein illustratedexemplary implementations of the invention. In addition, while aparticular feature may have been disclosed with respect to only one ofseveral implementations, such feature may be combined with one or moreother features of the other implementations as may be desired andadvantageous for any given or particular application.

What is claimed is:
 1. An apparatus of a mobile device for amplifyingradio frequency signals, comprising: a plurality of differentialamplifier cells coupled to an output passive network, wherein at leastone differential amplifier cell of the plurality of differentialamplifier cells comprises a pair of amplification transistors and a pairof cascode transistors, wherein the pair of amplification transistorscomprises a smaller gate width than the pair of cascode transistors; anddigital circuitry configurable to power on or off the at least one ofthe pair of cascode transistors.
 2. The apparatus of claim 1, whereinthe pair of cascode transistors are connected to drain terminals of thepair of amplification transistors, respectively.
 3. The apparatus ofclaim 1, wherein the output passive network comprises at least twoinductors.
 4. The apparatus of claim 1, wherein the pair ofamplification transistors comprise thin transistors and the pair ofcascode transistors comprise thick transistors that comprise a greatergate width than the thin transistors.
 5. The apparatus of claim 1,wherein the output passive network further comprises a singletransformer coupled to the plurality of differential amplifier cells. 6.The apparatus of claim 1, further comprising: a set of switches orinverters, coupled to the pair of amplification transistors, configuredto selectively enable a power level of the pair of amplificationtransistors via the digital circuitry based on a voltage threshold. 7.The apparatus of claim 1, wherein the output passive network isconfigured to enable a dual band operation as a multi-resonant matchingnetwork to enable a response for two different operating frequency bandsof the mobile device based on a single transformer connected in paralleldirectly to the plurality of differential amplifier cells.
 8. A systemof a mobile device for amplifying radio frequency signals, comprising: aplurality of differential amplifier cells coupled to an output passivenetwork, wherein at least one differential amplifier cell of theplurality of differential amplifier cells comprises a pair ofamplification transistors and a pair of cascode transistors, wherein thepair of cascode transistors comprises a wider gate than the pair ofamplification transistors; and digital circuitry configurable to operateat least one cascode transistor of the pair of cascode transistors. 9.The system of claim 8, wherein the pair of amplification transistors areconnected to source terminals of the pair of cascode transistors,respectively.
 10. The system of claim 8, wherein the output passivenetwork comprises a plurality of capacitive units and two inductorsforming a transformer.
 11. The system of claim 8, wherein the outputpassive network comprises a transformer connected in parallel to theplurality of differential amplifier cells.
 12. The system of claim 8,further comprising: a set of switches connected to plurality ofdifferential amplifier cells and configurable to enable a voltage inputto at least one of: the pair of amplification transistors or the pair ofcascode transistors.
 13. A method for a mobile device comprising:providing, via a plurality of differential amplifier cells coupled to anoutput passive network, power signals through a pair of amplificationtransistors of a differential amplifier cell of the plurality ofdifferential amplifier cells coupled to a pair of cascode transistors ofthe differential amplifier cell of the plurality of differentialamplifier cells that comprise a wider gate than the pair ofamplification transistors; and powering on or off, via digital circuitrycoupled to the plurality of differential amplifier cells, at least oneof the pair of cascode transistors.
 14. The apparatus of claim 13,wherein the pair of amplification transistors are connected to a driverterminal to receive a driver signal, and the pair of cascode transistorsare connected to a bias terminal to receive a bias signal.